Design and analysis of a multiple notched UWB-BPF based on microstrip-to-CPW transition

1 Introduction

Ultra wideband (UWB) systems mostly function in indoor or short-range environments due to the high data rate capability. However, these environments have other RF services (WLAN, C, X band, etc.) operational, and these narrowband RF pockets possess high power output within their specific domain. Although, UWB systems adhere to the Federal Communications Commission (FCC) output power level (−41.3 dBm/MHz) [ 1], but such crowded co-habitat of RF services generate multiple interference issues. To eliminate such unwanted scenarios, bandstop filters (BSFs) integrated to the UWB system becomes necessary. However, any addendum would lead to increase in circuit size. Hence, in this era of miniaturization, designers have kept circuit size minimum by integrating the BSF into the bandpass filter (BPF) via various methodologies [ 2], [ 3], [ 4], [ 5], [ 6], [ 7], [ 8], [ 9], [ 10], [ 11], [ 12], [ 13], [ 14], [ 15].

A very popular method of notch generation is with the use of stepped impedance resonator (SIR) along the microstrip [ 2], [ 3], [ 4] or input/output (I/O) lines [ 5]. The SIR in its original form [ 2], simplified composite right/left-handed (SCRLH) resonator form [ 3, 4], E-shaped resonator [ 6], or electromagnetic band gap (EBG) structures [ 7] has dual resonance properties which help in placing two TZs at points of interest in the passband. Folded microstrip lines utilize the principle of wave cancellation for generation of two TZs in [ 8, 9]. Complementary split ring resonators (CSRR) shaped slots and defected ground structures (DGS) develop dual notches in [ 10, 11] respectively. Multiple stubs attached to the microstrip generated dual notches in [ 12], whereas multiple passband TZs were developed in [ 13] using folded resonators placed in the slots of ground plane. Other dual notched band structures reported used the combination of any of the above mentioned principles [ 14, 15]. Folded lines combined with slots in microstrip were utilized for dual stopband generation in [ 14], whereas DGS with multiple SRRs placed two TZs in the passband of [ 15]. Most of the reported structures have been designed on high cost substrates [ 2], [ 3], [ 4], [ 5], [ 6, 8], [ 9], [ 10], [ 11], [ 12], [ 13], [ 14], [ 15], lack dual TZs at the spectrum edges resulting in poor frequency characteristics [ 2], [ 3], [ 4], [ 5], [ 6], [ 7, 10, 11, 14], [ 15], [ 16], have complex geometry due to use of vias [ 2], [ 3], [ 4], [ 5], [ 6], [ 7, 14, 16], and are of comparatively larger size [ 2], [ 3], [ 4], [ 5], [ 6], [ 7], [ 8], [ 9], [ 10], [ 11], [ 12], [ 13], [ 14], [ 15], [ 16].

In view of the above drawbacks we propose our structure. The basic design of this structure is based on broadband transition from microstrip-to-CPW, initially conceived in [ 17], and later exploited and applied to UWB-BPF generation in [ 18], [ 19], [ 20], [ 21]. The design essentially consists of a pair of microstrip-to-CPW transitions arranged back-to-back. This leads to generation of the quasi-elliptic basic UWB response with good insertion/return loss and dual TZs (flexible in positioning) at either passband edges. An analysis into the design of this basic UWB-BPF is provided. Later split ring resonators (SRRs) and folded SRRs (FSRRs) are placed close to the I/O feeding lines to develop the multiple in/out-band TZs. The TZs position and width can varied as per necessity. An approximate equivalent circuit model of the proposed dual-notched band is developed and verified against full wave EM simulation. The architecture of the proposed structure is depicted in Figure 1. The design and optimization of this structure has been carried out using the full wave EM software IE3D on the commercially viable FR4 substrate with ε_r = 4.4 and height 0.8 mm. The proposed structure is easy to design and fabricate due to the absence of vias. The proposed UWB filter is then fabricated to justify its predicted performance in S-magnitudes and group delay.

Figure 1:

The proposed microstrip-to-CPW transition based UWB filter. (a) Top plane, (b) ground plane. (c) Combined architecture. Dark shade conductor and white shade etched part. All dimensions in mm.

2 Proposed structure

The basic design of the proposed structure is a broadband balun that has two pairs of surface transitions on either sides of the substrate ( Figure 2a). Figure 2b and c respectively depict the equivalent and simplified equivalent circuits of the basic transition. In the equivalent circuit of Figure 2b, Z_0μs and Z_0CPW are the characteristics impedances of the microstrip and CPW respectively. The reactances of shorted slotline and open-circuited microstrip line are given by X_sl and X_μss respectively. The transformer ratio n represents the magnitude of coupling between microstrip on top and slotline in the ground, and is given by

where, Z_μss and Z_sl represent the impedances of microstrip stub line and slotline respectively at the coupling region. In the simplified equivalent circuit of Figure 2c, all the components have been transformed to the microstrip side. The input impedance on the left, looking into right is obtained from [ 21, 22] as:

for the open circuited microstrip stub X_μss = −Z_μss cot θ_μss, therefore

also,

and

whereas, for the shorted slotline,

Figure 2:

(a) Simplified transition of the proposed two pairs of microstrip-to-CPW. (b) Equivalent circuit of transitions and simplified equivalent circuit of the same.

Placing Eqs. (4)– (6) in Eq. (2), we get,

In order to design broadband transition, mutual cancellation of X_sl and X_μss is necessary, and this is achieved by placing θ_μss = θ_sl = π/2 in Eq. (7), which gives

For the basic balun, t_μss = 0.35 mm and w_sl = 0.2 mm give Z_μss = 98Ω and Z_sl = 90Ω respectively, i.e., from Eq. (1), n = Z_μss/Z_sl = 98/90 = 1.088 ≈ 1. The other optimized dimensions are w₀ = 1.5 mm (50Ω feedline), l_μss = 6.95 mm, l_sl = 4.65 mm, w_CPW = 1.5 mm, G = 5.84 mm, and g = 1.34 mm.

Subsequently, the CPW in ground ( Figure 3) is modified from uniform to non-uniform type (without any change to the microstrip on the top) so as to place its resonant modes with the designated UWB spectrum and develop an enhanced 3rd order quasi-elliptic response as portrayed in Figure 4. It is observed from Figure 4a that the 3-dB passband extends from 2.88-11 GHz with insertion loss better than 0.55 dB and return loss greater than 19.8 dB.

Figure 3:

Comparative frequency responses during the development (inset: sequential development of the ground plane).

Figure 4:

(a) Optimized response of the four-pole basic UWB-BPF. Current distributions at respective frequencies of (b) 4.8 GHz, (c) 6.5 GHz and (d) 10.3 GHz.

Current distributions at respective resonant frequencies 4.8 GHz, 6.5 and 10.3 GHz are depicted in Figure 4b–d. Two TZs at 1.1 and 11.5 GHz, with attenuations greater than 70 and 30 dB respectively, bring about enhanced selectivity. These two TZs are functions of the wider slot dimensions (a, b) of the CPW and their variability is depicted in Figure 5. The stopband extends beyond 15 GHz with attenuation greater than 10 dB. The optimized dimensions of the basic UWB filter are, a = 4.5 mm, b = 3.8 mm, w₀ = 1.5 mm, x₁ = 4.5 mm, y₁ = 1.5 mm, x₂ = 6.84 mm, y₂ = 1.0 mm, p = 2.355 mm, g = 1.14 mm, l_μSS = 6.91 mm_, t_μSS = 0.35 mm_, G = 6.14 mm (≈λ_gmicrostrip/4), L = 21.6 mm, W = 16 mm.

Figure 5:

(a) Variation in upper cut-off frequency due to changes in a. (b) Variation in lower cut-off frequency due to many changes in b.

3 Multiple notched bands development

The UWB spectrum is a crowded environment of multiple in-band RF sources like WLAN, C, X bands etc., which regularly act as sources of interferences. Hence, for the present UWB filter we intend to place multiple TZs within the UWB passband spectrum at 6 and 8.1 GHz so as to mitigate in-band interference from WLAN and X band respectively. In order to achieve the same, multiple FSRRs and SRRs are placed adjacent to the I/O feeding lines respectively. The FSRR has electrical (and physical) length greater than the SRR and hence used for generating TZs deep within the passband. Figure 6a shows the variable positioning of TZs for different lengths of the FSRR and SRR. The lengths of FSRR and SRR are related to their respective notch frequencies as:

Figure 6:

(a) Variable dual notch positions for different lengths of FSRR and SRR. (b) Variation in notch width. (c) Proposed dual notched band UWB-BPF for optimized dimensions of the structure.

These FSRRs and SRRs when optimized in length (l_FSRR = 16.8 mm and l_SRR = 12.2 mm), generate TZs at the frequency of interest (6 and 8.1 GHz respectively) within the passband. Also, the notch width can be manipulated by embedding similar multiple resonators as depicted in Figure 6b. Table 1 provides the 3 and 10 dB fractional bandwidths for multiple resonators. The optimized dual notched frequency response of the proposed structure is presented in Figure 6c from which it can be observed that the passband edges have TZs at 1.1 and 11.45 GHz respectively. The UWB spectrum extends from 2.8–10.8 GHz with two passband TZs at 6 and 8.1 GHz. These passband TZs have attenuations greater than 26 and 21 dB respectively. The passband insertion/return loss is better than 0.45/21 dB before the first notch, 0.38/17 dB between the two notches and 0.28/15 dB after the second notch. The stopband is suppressed till 15.5 GHz with attenuation better than 18 dB.

The presence of dual notches makes the UWB response look like tri-band BPF response [ 23].

Surface current distribution of the notched band filter structure is shown in Figure 7, which depicts the coupling (activeness) of the resonators (FSRR and SRR) at their respective resonant frequencies. From Figure 7a and b it is observed that at 6 and 8.1 GHz, FSRR and SRR respectively have very high concentration of current. This means that at its resonant frequency (6 GHz), the FSRR is strongly coupled to the basic UWB filter and at 8.1 GHz, SRR is strongly coupled to the BPF. Moreover, Figure 7a and b also gives us the information that the direction of currents in the resonators and the stubs of BPF are in opposite directions, at their respective resonant frequency, which leads to creation of transmission zeros (TZs) due to the wave cancellation principle.

Figure 7:

(a) Current distribution at 6 GHz and (b) current distribution at 8.1 GHz.

An approximate lumped equivalent circuit model of the proposed band notched UWB filter is constructed, as observed in Figure 8a. The figure depicts the representation of various distributed components of the proposed structure by their lumped counterpart. The feed lines are represented by components L₀, L₁ and C₁. The stubs appended to the feedlines are depicted by parallel combination of L₂, C₂ and are coupled to each other via C₀. These stubs are capacitively coupled to the CPW in ground through C₈ and C₉, whereas the CPW is portrayed by the tank circuit of L₃, C₃ and C₁₀. The FSRR and SRR are represented by parallel combination of L₄, C₄ and L₅, C₅ respectively, and are coupled to the stubs/CPW via C₇ and C₆. The dual notch positions are related to the lumped parameters through formulas given below.

Figure 8:

(a) Approximate equivalent circuit of the proposed structure. (b) Comparative S parameters responses of full wave EM and circuit simulation.

The optimized lumped parameters are L₀ = 0.55 nH, L₁ = 1.06 nH, L₂ = 3.06 nH, L₃ = 0.733 nH, L₄ = 1.312 nH, L₅ = 1.01 nH, and C₀ = 0.0047 pF, C₁ = 0.114 pF, C₂ = 4.2 pF, C₃ = 0.497 pF, C₄ = 0.175 pF, C₅ = 0.374 pF, C₆ = 0.3286 pF, C₇ = 0.118 pF, C₈ = 0.22 pF, C₉ = 0.22 pF, C₁₀ = 0.4226 pF. The comparative response of the circuit simulation and full wave EM simulation is plotted in Figure 8b.

4 Experimental verification

A prototype of the proposed structure is developed and its measured results are verified against the simulated results ( Figure 9). The measurement is done using Vector Network Analyzer N5230A. The response demonstrates a passband bandwidth from 2.9–10.7 GHz, which covers the UWB spectrum specified by the FCC. The dual notches are positioned at 6 and 8.05 GHz with attenuation greater than 18 dB. The insertion loss throughout the passband (except the dual notches) is better than 1.3 dB and return loss greater than 12 dB. The stopband is wide till 16 GHz with attenuation larger than 17 dB. The group delay observed is small and flat varying within the range of 0.11–0.3 ns. A comparative study of the proposed structure is carried out against those available in literature in Table 2. It can see from column 2 of the table that the proposed structure meets the UWB passband requirement comfortably. The measured insertion loss of the proposed structure is better than [ 12, 14], comparable with [ 13], and subpar with others. The subpar performance maybe due to human error involved in fabrication. The data on insertion loss of [ 3], [ 4], [ 5], [ 6], [ 7, 10] is not provided in their respective literature.

Figure 9:

Comparative frequency characteristics of measured and simulated data.

However, in simulation, the insertion loss of basic UWB filter as well as that of the proposed dual notched band UWB filter are better than (or at least comparable to) most [ 2], [ 3], [ 4], [ 5], [ 6], [ 7], [ 8], [ 9], [ 10], [ 11], [ 12], [ 13], [ 14], [ 15], [ 16]. Column 4 depicts that the position of measured dual notches are very close to the simulated ones and so are their attenuation levels. The proposed structure has stopband performance better than [ 3, 4, 6], [ 7], [ 8, 12, 13, 16], comparable with [ 15] and below that of [ 5, 9], [ 10], [ 11, 14]. However, its compulsory that better stopband performance should not come at the compensation of poor passband performance. The last column sheds light on the compactness of the proposed structure. The proposed selective UWB filter is simple to design using a commercial full wave EM simulation software and relatively easy to fabricate using conventional wet chemical etching method due to absence of vias (which requires drilling and soldering). The filter structures mentioned in [ 2], [ 3], [ 4], [ 5], [ 6], [ 7, 14, 16] use via which calls for professional expertise in drilling and soldering as the substrates are usually thin and additional drilling and soldering efforts may affect the performance of the filter adversely due to potential impedance variations which cause impedance mismatch at those weak links if not accomplished in perfect manner or may even break the structure. The dual notched structures of [ 8, 9] are simple to design but because the spacing between arms of the resonators and feedlines are 0.05 mm, it is usually difficult to realize these dimensions using conventional wet chemical etching method accurately. Also, the proposed BPF possesses TZs with good roll-off at both the passband edges better than [ 2], [ 3], [ 4], [ 5], [ 6], [ 7, 10, 11, 14], [ 15], [ 16] and is physically of size smaller than [ 2], [ 3], [ 4], [ 5], [ 6], [ 7], [ 8], [ 9], [ 10], [ 11, 14, 16]. Considering all these issues mentioned above, authors think their reported planar compact selective UWB filter is relatively simple in topology and has frequency characteristics better or that most dual notched structures reported in literature.

5 Conclusions

The manuscript reports a dual notched band UWB-BPF with quasi-elliptic response on a low cost substrate. The basic 3rd order UWB filter is constructed on technology of broadband balun of surface-to-surface coupled structure, here, microstrip on top with CPW in ground. This topology displays multiple TZs at the passband edges and within the stopband which brings about good insertion/return loss, sharp roll-off, minimum group delay and extended stopband respectively. Dual notches are placed at points of interest within the passband via wave-cancellation technique using SRRs and FSRRs coupled to I/O feed lines. The developed prototype is of compact nature, hence can be easily integrated with modern day communication systems.